Technical Field
The present disclosure relates to controlling multiphase power factor correction (PFC) converters and, in particular, adaptively interleaving a plurality of multiphase PFC converters.
Description of the Related Art
It is generally known to use PFC devices for actively correcting the power factor of switched-mode power supplies (SMPS) used to supply electronic apparatuses, such as computers, televisions, monitors, etc., and lighting devices such as fluorescent lamps. A typical SMPS comprises a full-wave diode rectifier bridge, having an input connected to the AC power distribution line, and a capacitor connected downstream so as to produce a DC voltage from the AC supply voltage. The capacitor has a large enough capacitance for a relatively small ripple to be present at its terminals as compared to a DC level. Therefore the rectifier diodes of the bridge will only conduct over a short portion of each half cycle of the supply voltage, as the instantaneous value thereof is less than the voltage of the capacitor over most of the cycle. The result is the current absorbed from the power line consists of a series of short impulses the amplitude of which is 5-10 times the resulting average value.
This has significant consequences:
the current absorbed from the power line has peak and rms (root-mean-square) values much higher than the case of sinusoidal current absorption,
the supply voltage is distorted due to the almost simultaneous impulse absorption of all utilities connected to the power line,
the current in the neutral conductor in the case of three-phase systems is highly increased, and
there is low use of the energy potential of the power system.
In fact, the waveform of impulse current includes many odd harmonics, which although do not contribute to the power provided to the load, they contribute to increasing the rms current absorbed by the power line and, therefore, to increasing the energy dissipation.
In quantitative terms, this may be expressed both in terms of power factor (PF), which is a ratio of the real power (the one the power supply sends to the load plus the one dissipated therein in the form of heat) to the apparent power (the product of the rms voltage by the rms current absorbed), and in terms of total harmonic distortion (THD), generally a percentage ratio of the energy associated with all larger harmonics to the one associated with the fundamental harmonic. Typically, a power supply with capacitance filter has a PF between 0.4 and 0.6 and a THD higher than 100%. A PFC arranged between the rectifier bridge and output allows a current quasi sinusoidal and phased with the voltage, to be absorbed from the network, thus making the PF close to 1 and decreasing the THD.
FIG. 1 shows a schematic of a single phase PFC converter 20 and a control device 22. The control device 22 may have a variable frequency and may operate between the continuous (CCM) and discontinuous (DCM) modes in what is commonly called “Transition Mode” (TM). An input voltage (Vin) is supplied to the converter 20 from a voltage supply terminal of a full-wave diode rectifier bridge 24. A second terminal of the rectifier bridge 24 is, in turn, connected to a ground terminal. The rectifier bridge 24 receives an input supply voltage provided by an alternating current (AC) power supply 26. The converter 20 may be implemented as a boost converter. The converter 20 comprises an inductance 28 of a transformer, a metal oxide semiconductor (MOS) power transistor 30, and a diode 32. A first terminal of the inductance 28 is connected to the voltage supply terminal of the rectifier bridge 24 and the drain terminal of the transistor 30 is connected to a second terminal of the inductance 28 downstream from the first terminal. The source terminal of the transistor 30, on the other hand, is connected to a ground terminal via a resistor 34.
The diode 32 has an anode connected to the second terminal of the inductance 28. Further, the diode 32 has a cathode connected to a first terminal of an output capacitor 36 having another terminal connected to a ground terminal. The converter 20 generates a direct current (DC) output voltage (denoted as Vout) across the output capacitor 36, whereby Vout is higher than the maximum peak voltage supply of Vin.
The control device 22 keeps the output voltage Vout at a constant value using feedback control. In transition mode, efficient switching is achieved by switching at zero voltage and zero current conditions. The control device 22 of FIG. 1 is shown to have four input ports and an output port. The four input ports include an input voltage port 38 that receives the input voltage provided by the rectifier bridge 24, an output voltage port 40 that receives the output voltage provided by the converter 20, a zero crossing detection (ZCD) port 42, and current sense (CS) port 44. The ZCD port 42 is electrically coupled to an auxiliary winding 46 of the transformer that includes the inductance 28 and is for sensing when the current through the inductance 28 reaches 0 Amperes. The CS port 44 is used to monitor the current through the transistor 30.
The output port 48 of the control device 22, which is a gate drive (GD) port, turns the transistor 30 on and off and thereby controls the operation of the converter 20. The control device 22 may be said to be a variable frequency control device because the both the frequency of switching the transistor 30, and, accordingly, the switching period, are based on external events triggered by the operation of the converter 20. Operational efficiency is achieved, however, by ensuring that the transistor 30 is turned on at near zero current of the inductance 28.
Furthermore, the control device 22 may be of a constant on-time (COT) type. In constant on-time operation, the turn-on period of the power transistor 30 of the converter 20 is used as a control variable and, during each cycle of voltage supply, it is kept constant at an appropriate value to obtain the regulation of the voltage output from the converter 20.
FIG. 2 shows timing diagrams of the signals of the circuit of FIG. 1 when Vin is less than half of Vout and greater than half of Vout. As may be viewed on the left side of the Figure, when Vin is less than half of Vout the transistor 30 is turned on for more than half of the length of the switching period, Tsw. The transistor 30 is turned off for the remainder of the switching period, however, by setting VGS (the gate voltage) to zero. The transistor 30 is turned back on again after a zero crossing condition is met (or ZCD voltage dropping below a threshold). Similar operation is shown on the right side of FIG. 2, where the on-time of the transistor 30 is shorter because the desired output voltage is less.
The converter 20 is characterized by a high current ripple through the inductance 28 and the input and output terminals. The current ripple may be significantly reduced in multiphase parallel converters where two converters, which may be implemented as a boost converters, are connected in parallel to provide an output voltage. The reduction of the current ripples results from operating the two parallel converters out of phase with respect to each other, whereby the ripples from the converters cancel out one another. The maximum reduction occurs when the boost converters are operated at a 180 degree phase difference.
FIG. 3 shows a schematic of an overall controller 60 comprising two converters 64a,b and a multiphase control device 62. Although two converters 64a,b are shown in FIG. 3, it is noted that a different number of converters may be used. Further, the converters 64a,b are similar to the converter 20 described with reference to FIG. 1, whereby each converter 64a,b may be a boost converter. The converters 64a,b are supplied with voltage via a rectifier bridge 24 connected to an AC power supply 26. The converters 64a,b are similarly configured as those of the converter 20, whereby each converter 64a,b comprises an inductance 66a,b of a transformer, a transistor 68a,b, a diode 70a,b, and a resistor 72a,b, respectively. Each transformer has an auxiliary winding 65a,b using which zero crossing conditions are detected.
Furthermore, the converters 64a,b are connected in parallel to both the DC voltage (Vin) output by the rectifier bridge 24 and to an output capacitor 36 across which the output voltage (Vout) is obtained.
The multiphase control device 62 controls the operation of the converters 64a,b by timing the turning on of the transistors 68a,b through gate drive ports 74a,b, respectively. It is noted that as an alternative to using one multiphase control device 62 to control both converters 64a,b multiple single phase control devices may be used for turning on the transistors 68a,b. 
Conventional approaches to keeping the converters 64a,b completely out of sync and operating 180 degrees apart include allowing one converter (for example, first converter 64a) to operate as if it was a single phase converter. The first transistor 68a of the first converter 64a is switched on when zero crossing conditions of the inductance 66a are detected and switched off at a later time period.
Meanwhile, the second converter 64b is turned on at half of the switching period of the first converter 64a (i.e., half a cycle following switching on the first transistor 68a) independently of whether the second converter 64b has reached the zero crossing conditions. However, this approach has drawbacks because the second transistor 68b may be turned on when the current is not zero. Furthermore, it was observed that this approach may make certain conditions unstable.
Another conventional approach calls for utilizing a phase-locked loop (PLL) to keep the converters 64a,b synchronized to a 180 degree phase difference. A control device is used to detect the turn on instant of the converters 64a,b and if they are not 180 degrees apart, the PLL changes the on-time of the converters 64a,b to bring them back in phase. However, the PLL requires some time to lock particularly because the converters 64a,b are continually changing frequency.